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\begin{document}

\title{Implementation of a novel buck-boost converter based on coupled inductor
for renewable energy applications}



\author[1]{Peng Luo}%
\author[1]{Yongyan Chen}%
\author[1]{Junzhe Hong}%
\author[1]{Lanyi Zhou}%
\author[1]{Haoyu Jiang}%
\affil[1]{Guangdong Ocean University}%


\vspace{-1em}



  
  \date{May 17, 2024}


\begingroup
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\selectlanguage{english}
\begin{abstract}
A coupled inductor based on dual switches buck-boost converter is
proposed in the article, which can be used in renewable energy
applications. Wide conversion ratio can be realized by regulating turns
ratio of the coupled inductor and the duty cycle. The topology has a
quadratic- like voltage gain, and the low voltage stress and current
stress. Combing the passive clamping circuit to recover leakage energy,
the voltage spikes of power switches are suppressed, improving the
efficiency. Moreover, it has the merits of continuous input current and
common ground. To demonstrate the superior performances of the presented
converter, the principle of operation and steady-state analysis are
presented in detail. At last, a 100 W experimental prototype was
designed to verify the correctness of the theoretical analysis, with a
maximum efficiency of 95.7\% in step-up mode and 96.8 \% in step-down
mode.%
\end{abstract}%



\sloppy


\emph{ORIGINAL PAPER}

\textbf{Implementation of a novel buck-boost converter based on coupled
inductor for renewable energy applications}

Peng Luo \textbar{} Yongyan Chen \textbar{} Junzhe Hong \textbar{} Lanyi
Zhou \textbar{} Haoyu Jiang*

School of Electronic and Information Engineering, Guangdong Ocean
University, Zhanjiang, China

\textbf{Correspondence}

*Haoyu Jiang, School of Electronic and Information Engineering,
Guangdong Ocean University, Zhanjiang, China. Email:
john\_h\_y\_walter@163.com

\textbf{Abstract}

A coupled inductor based on dual switches buck-boost converter is
proposed in the article, which can be used in renewable energy
applications. Wide conversion ratio can be realized by regulating turns
ratio of the coupled inductor and the duty cycle. The topology has a
quadratic- like voltage gain, and the low voltage stress and current
stress. Combing the passive clamping circuit to recover leakage energy,
the voltage spikes of power switches are suppressed, improving the
efficiency. Moreover, it has the merits of continuous input current and
common ground. To demonstrate the superior performances of the presented
converter, the principle of operation and steady-state analysis are
presented in detail. At last, a 100 W experimental prototype was
designed to verify the correctness of the theoretical analysis, with a
maximum efficiency of 95.7\% in step-up mode and 96.8 \% in step-down
mode.

\textbf{KEYWORDS}

Buck-boost converter, coupled inductor, renewable energy

\section*{INTRODUCTION}

{\label{introduction}}

Due to the scarcity of the improvement of environmental awareness and
fossil fuel energy, the renewable energy has drawn more
attention\textsuperscript{1,2}. Photovoltaic (PV) and wind power play a
crucial role in industrial applications, reducing the
pollution\textsuperscript{3}. However, climate and weather patterns can
cause renewable energy sources to produce inconsistent and intermittent
output voltage \textsuperscript{4}. Buck-boost converter can regulate
the voltage, which is suitable for renewable energy
conversion\textsuperscript{5}.

The typical buck-boost converter can achieve wide range of input voltage
with simple structure, which has been researched in recent
years\textsuperscript{6,7}. However, its applications are limited by
voltage gain and non-common ground\textsuperscript{8,9}. To achieve
these issues, the CUK, ZETA and SEPIC converter basic on buck-boost
converter are proposed. However, the semiconductors are withstand high
voltage stress and current stress \textsuperscript{10}.

To improve the above shortcomings, some transformerless single switch
buck-boost converters are proposed. In Banaei and
Bonab\textsuperscript{11}, a ZETA converter based buck-boost circuit is
presented to achieve higher voltage gain than traditional buck-boost
converter. However, the improvement of voltage gain is limited.
Converters in Gorji, et al.\textsuperscript{12} and Shu, et
al.\textsuperscript{13} propose the quadratic voltage gain buck-boost
converter that effectively increase the voltage with the narrow range of
duty cycle. But the above quadratic buck-boost converters have
disadvantage of high input current ripple. To achieve continuous input
current and decrease the input ripple, the inductor is joined
straightforwardly to the input source.\textsuperscript{14,15}. Combing
the quadratic cell and continues input current technology, converter in
Zhang, et al.\textsuperscript{16} and Kumar and
Krishnasamy\textsuperscript{17} are proposed to obtain high voltage gain
only by adjusting the duty cycle. Coupled inductor is another technology
which has been used widely in converter to improve the voltage gain by
regulating the turns ratio. Considering the leakage inductor of coupled
inductor, the passive clamp circuit can be employed to recycle the
energy of leakage inductor and suppress the voltage spike of power
switch\textsuperscript{18,19}. Rong, et al.\textsuperscript{20}proposes
a passive clamp circuit-based buck-boost-Cuk converter that employs a
coupled inductor to achieve higher voltage gain. But it is a non-common
ground structure which increases the difficulty of control system and
electromagnetic interference (EMI). In Alizadeh, et
al.\textsuperscript{21}, a single switch quadratic buck-boost converter
based on coupled inductor is presented, which has the benefits of common
ground and continuous input current. Also, it has the disadvantage that
the voltage stress of the power switch is high.

To decrease the voltage stress and current stress on the semiconductor,
dual switches structure is widely utilizing in power electronics
converter\textsuperscript{22,23}. Some dual switches buck-boost
converters are proposed in the recent time. Combing continuous input and
quadratic, a dual switches converter is proposed\textsuperscript{24}. To
further decrease the voltage stress of the power switches, a family of
cascading of boost and ZETA structure converters are proposed in
Veerachary and Khuntia\textsuperscript{25}. Compared with the above
relevant dual switches converter, converter in Okati, et
al.\textsuperscript{26} has higher conversion ratio. To solve the effect
of non-common and hold on the voltage gain with previous converter, a
modified SEPIC converter is proposed in Wang, et
al.\textsuperscript{27}. A novel negative output buck-boost converter in
Ding and Wang\textsuperscript{28} and a converter with interleaving two
CUK converters in Taghizadegan Kalantari, et al.\textsuperscript{29} are
proposed, which achieve wider conversion ratio than above dual switches
converter and have non-common ground structure. Converter in
Hosseinpour, et al.\textsuperscript{30} proposes a dual switches
converter with common ground structure. The circuit achieves high
voltage gain while minimizing the voltage stress of the switch through
increment passive components' number, which make the converter bulky.

Considering the above technology, this article proposes a dual switches
buck-boost converter based on coupled inductor. Applying coupled
inductor, high voltage gain can be attained by regulating both the duty
cycle and the turns ratio. The quadratic- like voltage gain is achieved
to wide conversion ratio. Moreover, the voltage stress and current
stress are lower than traditional quadratic cell. The passive clamp
circuit effectively suppresses voltage spike of the power switches, also
recycling the energy of leakage inductor. Common ground between input
sources and load reduces the EMI and difficulty of the control design.
Therefore, the proposed converter can attain output voltage with
stability and suitability for solving the issues of renewable energy
applications.

In this paper, Section 2 analyzes the operation principles. The
stead-state analysis, which includes the voltage gain and stresses of
components, is discussed in Section 3. The component parameter design
and power loss analysis for the proposed converter are discussed in
Section 4 and Section 5, respectively. Section 6 is a comparison of the
characteristics of the relevant converters which can demonstrate the
superior of the presented converter. Section 7 demonstrates the
experiment results with open loop and close loop in both step-up and
step-down modes.

\section*{OPERATION PRINCIPLES}

{\label{operation-principles}}

The equivalent circuit of dual switches coupled inductor buck-boost
converter is given in Fig. 1, which is assembled from two power switches
\emph{S} \textsubscript{1} and\emph{S} \textsubscript{2}, three
capacitors \emph{C} \textsubscript{1},\emph{C} \textsubscript{2}, and
\emph{C\textsubscript{o}} , three diodes\emph{D} \textsubscript{1},
\emph{D} \textsubscript{2} and\emph{D} \textsubscript{3}, an inductor
\emph{L} \textsubscript{1}, and a coupled inductor. Coupled inductor
combines with leakage inductor\emph{L\textsubscript{k}} , magnetizing
inductor\emph{L\textsubscript{m}} and ideal transformer with primary
winding\emph{L\textsubscript{p}} and secondary
winding\emph{L\textsubscript{s}} . Two power switches operate
synchronously. To simplify the calculation of steady-state analysis,
several assumptions are considered as follow:

\begin{enumerate}
\tightlist
\item
  The values of capacitors are large enough so that the voltage of each
  capacitor is regarded as constant in a switching cycle.
\item
  The parasitic resistance and forward voltage drop of the components
  can be ignored.
\item
  The turns ratio of coupled inductor can be defined
  as\emph{N\textsubscript{p}} : \emph{N\textsubscript{s}} = 1 : n.
\item
  The coupling coefficient \emph{k} is defined as
  \emph{L\textsubscript{m}} / (\emph{L\textsubscript{m}} +
  \emph{L\textsubscript{k}} ).
\end{enumerate}

Fig. Equivalent circuit.

Fig. 2 shows the voltage and current waveforms with main components of
the converter which operates in continuous conduction mode (CCM).
Current flow paths of three operating modes in a switching cycle are
plotted in Fig. 3.

Mode I (\emph{t} \textsubscript{0}-\emph{t} \textsubscript{1}) {[}Fig.
3(a){]}: The power switches \emph{S} \textsubscript{1} and\emph{S}
\textsubscript{2} are turned on; diodes\emph{D} \textsubscript{1} and
\emph{D} \textsubscript{3} are reverse biased, diode \emph{D}
\textsubscript{2} is on. The magnetizing
inductor\emph{L\textsubscript{m}} is charged by
\emph{V\textsubscript{in}} , and capacitors \emph{C} \textsubscript{1}
and \emph{C} \textsubscript{2}release energy to inductor \emph{L}
\textsubscript{1}.\emph{C\textsubscript{o}} provides energy to the
load.\emph{i\textsubscript{L}} \textsubscript{1} starts to increase
and\emph{i\textsubscript{Lk}} increases rapidly. This mode ends
while\emph{D} \textsubscript{2} is off.

Mode II (\emph{t} \textsubscript{1}-\emph{t} \textsubscript{2}) {[}Fig.
3(b){]}: The power switches \emph{S} \textsubscript{1} and\emph{S}
\textsubscript{2} are still remain activated while all diodes are
reverse biased. Other current flow paths are same as Mode I. Besides,
the currents of \emph{L\textsubscript{m}} and\emph{L} \textsubscript{1}
increase. \emph{i\textsubscript{Lm}} starts to increase. When the signal
of power switches becomes low, this mode ends. Some equations can be
obtained in this mode as following.

()

()

()

()

Mode III (\emph{t} \textsubscript{2}-\emph{t} \textsubscript{3}) {[}Fig.
3(c){]}: The power switches \emph{S} \textsubscript{1} and\emph{S}
\textsubscript{2} are turned off simultaneously. Diodes\emph{D}
\textsubscript{1}, \emph{D} \textsubscript{2}, and\emph{D}
\textsubscript{3} are forward biased. Capacitor\emph{C}
\textsubscript{1} is charged by the magnetizing
inductor\emph{L\textsubscript{m}} and \emph{V\textsubscript{in}} .
Meanwhile, the magnetizing inductor \emph{L\textsubscript{m}} transfers
energy to the secondary side of coupled inductor. Capacitor \emph{C}
\textsubscript{2}stores the energy through the coupled inductor
and\emph{D} \textsubscript{2}. The inductor \emph{L}
\textsubscript{1}releases energy to the output capacitor
\emph{C\textsubscript{o}} and the load through \emph{D}
\textsubscript{3}. The current values of\emph{L\textsubscript{k}} ,
\emph{L\textsubscript{m}} and\emph{L} \textsubscript{1} are above zero
and decrease. When the signal of power switches becomes high, this mode
ends. Some equations can be obtained in this mode as follows:

()

()

()

()

()

(a) (b)

Fig. The key waveforms of the main components under (a) step-up mode;
(b) step-down mode.

(a)

(b)

(c)

Fig. . The operating modes (a) Mode I, (b) Mode II, (c) Mode III.

\section*{STEADY-STATE ANALYSIS}

{\label{steady-state-analysis}}

\subsection*{Voltage Gain}

{\label{voltage-gain}}

Utilizing (2) and (6), by applying the principle of voltage-second
balance on the magnetizing inductor \emph{Lm} , (10) can be deduced as
following.

()

The voltage across \emph{C} \textsubscript{1} can be given.

()

Using (6) and (11), the voltage stress of \emph{C} \textsubscript{2} can
be obtained.

()

From (1) and (5), by applying the principle of voltage-second balance
on\emph{L} \textsubscript{1}, the voltage gain \emph{M} can be achieved
as

()

Substituting (11) and (12) into (13), the voltage gain \emph{M} can be
achieved.

()

The correlation among voltage gain \emph{M} , duty cycle \emph{D} , and
coupling coefficient \emph{k} under \emph{n} =1 is illustrated in Fig.
4. The coupling coefficient has almost no impacts to the voltage gain.
Hence the coupling coefficient can be setting as \emph{k} =1. The ideal
voltage gain can be calculated in (15).

()

Fig. . The correlation among voltage gain \emph{M} , duty cycle \emph{D}
, and coupling coefficient\emph{k} .

\subsection*{Semiconductors' Voltage
Stresses}

{\label{semiconductors-voltage-stresses}}

When power switches are turned on, using Kirchhoff's Voltage Law (KVL),
from (1), (11), and (12), the voltage stresses of diodes are derived as

()

()

()

From Fig. 3 (c), (11), (12), and (15), the voltage stresses of power
switches are expressed as

()

()

\subsection*{Semiconductors' Current
Stresses}

{\label{semiconductors-current-stresses}}

During a switching cycle, applying the Kirchhoff's Current Law (KCL),
the currents of the capacitors \emph{C\textsubscript{o}} and\emph{C}
\textsubscript{1} can be written.

()

()

By applying the principle of ampere-second balance
capacitor\emph{C\textsubscript{o}} , the average current of the
inductor\emph{L} \textsubscript{1} can be achieved as (23).

()

Similarly, based on (22), the average current across\emph{D}
\textsubscript{1} in conducting time is presented as

()

From (24), during a switching cycle, we can derive the average current
of \emph{D} \textsubscript{1} as

()

From Fig. 1 and (23), according to KCL and the principle of
ampere-second balance, the following average currents of\emph{D}
\textsubscript{3} and \emph{D} \textsubscript{2} can be given,
respectively.

()

()

Similarly, using (15) and (27), the average currents flowing through the
power switches \emph{S} \textsubscript{1} and \emph{S}
\textsubscript{2}can be written as follows:

()

()

Applying (27) and (28), the average current flowing through the
magnetizing inductor \emph{L\textsubscript{m}} is given similarly

()

Calculating the current ripples of inductors \emph{L}
\textsubscript{1}and \emph{L\textsubscript{m}} can be achieved as (31)
and (32), respectively.

()

()

According to (23), (30)-(32), the maximum and minimum currents
of\emph{L} \textsubscript{1} and \emph{L\textsubscript{m}} are achieved.

()

()

()

()

According to KCL, the currents of diodes \emph{D}
\textsubscript{1},\emph{D} \textsubscript{2} and \emph{D}
\textsubscript{3} during a switching cycle can be written as follows:

()

()

()

Utilizing (37)-(39), the root-mean-square (RMS) currents of the diodes
can be derived.

()

()

()

Within one switching cycle, the expressions of the leakage inductor
current \emph{i\textsubscript{Lk}} and the secondary winding
current\emph{i\textsubscript{Ls}} through the coupling inductor are as
follows, respectively.

()

()

According to the (23), (43) and (44), the RMS currents of the magnetics
components can be derived.

()

()

()

In a switching cycle, the currents of capacitors\emph{C}
\textsubscript{1}, \emph{C} \textsubscript{2}
and\emph{C\textsubscript{o}} can be derived applying KCL.

()

()

()

From (48)-(50), the RMS currents flowing through the capacitors are
expressed as

()

()

()

During a switching cycle, according to KCL, the currents of the power
switches can be derived as.

()

()

From the equations (54) and (55), the RMS currents flowing through the
power switches can be deduced.

()

()

Due to the power switches are on during Mode II, the peak currents
flowing through the power switches can be presented as

()

()

For the power switches, the current expressions for these two
instantaneous moments can be derived.

()

()

\section*{COMPONENT PARAMETER DESIGN}

{\label{component-parameter-design}}

\subsection*{Design of Inductors}

{\label{design-of-inductors}}

Assuming that \emph{I\textsubscript{oB}} is the output current of
boundary conduction mode (BCM). The average currents flowing
through\emph{L} \textsubscript{1} and \emph{L\textsubscript{m}} in BCM
are equal to half of the current ripple of \emph{L} \textsubscript{1}
and\emph{L\textsubscript{m}} , respectively.

()

()

According to (62) and (63), the values of inductors\emph{L}
\textsubscript{1} and \emph{L\textsubscript{m}} can be deduced.

()

()

\subsection*{Design of Capacitors}

{\label{design-of-capacitors}}

Considering the voltage ripples of capacitors, the values of capacitor
are designed suitable to increase the efficiency and stability. The
voltage ripples \selectlanguage{greek}Δ\selectlanguage{english}\emph{v\textsubscript{C1}} ,
\selectlanguage{greek}Δ\selectlanguage{english}\emph{v\textsubscript{C2}}and \selectlanguage{greek}Δ\selectlanguage{english}\emph{v\textsubscript{Co}} can be
derived as follows:

()

()

()

\section*{POWER LOSS ANALYSIS}

{\label{power-loss-analysis}}

To analyze the presented converter efficiency, the power loss is divided
into four parts, including inductors loss, power switches loss, diodes
loss, capacitors loss. The parasitic parameters are defined as
follows:\emph{r\textsubscript{L}} \textsubscript{1},
\emph{r\textsubscript{Lp}} , and \emph{r\textsubscript{Ls}} are the
equivalent series resistance (ESR) of the inductor \emph{L}
\textsubscript{1} and the winding\emph{L\textsubscript{p}} and
\emph{L\textsubscript{s}} , respectively;\emph{P\textsubscript{cv,L}}
\textsubscript{1} and\emph{P\textsubscript{cv,cp}} are the core losses
per unit volume, and\emph{V\textsubscript{e,L}} \textsubscript{1}
and\emph{V\textsubscript{e,cp}} are the core volumes of inductor\emph{L}
\textsubscript{1} and coupled-inductor,
respectively;\emph{r\textsubscript{ds}} is the power switch's
on-resistance;\emph{t\textsubscript{on}} is the summary of rise time and
delay time and \emph{t\textsubscript{off}} is the summary of fall time
and delay time; \emph{V\textsubscript{Dx}} and
\emph{r\textsubscript{Dx}} are the diodes forward drop voltage and
forward resistance of diodes\emph{D\textsubscript{x}} (\emph{x} =1, 2,
3)\textsubscript{,}respectively; \emph{r\textsubscript{Ci}} is the ESR
of the capacitors\emph{C\textsubscript{i}} (\emph{i} = 1, 2, \emph{o} ).
The power loss analysis in detail will be expressed as following.

\subsection*{Power Loss of Inductors}

{\label{power-loss-of-inductors}}

Inductors power loss is divided into winding loss and core loss. The
winding loss relies on the flowing current and ESR of winding, which can
be derived as

()

The core losses of inductor \emph{L} \textsubscript{1} and coupled
inductor can be calculated as

()

From (69) and (70), the total loss of the inductors can be expressed as

()

\subsection*{Power Loss of Power
Switches}

{\label{power-loss-of-power-switches}}

The power loss of power switches includes the conduction loss caused by
on-resistance, and the switching loss caused by the transition times
when the power switches are turning on and turning off.

The power switches' conduction and switching losses of the power
switches can be calculated as (72) and (73).

()

()

Therefore, (74) gives the total loss of \emph{S} \textsubscript{1}
and\emph{S} \textsubscript{2}.

()

\subsection*{Power Loss of Diodes}

{\label{power-loss-of-diodes}}

The power loss of the diodes is mainly caused by the forward voltage
drops and resistance of diodes, which can be derive as

()

()

Therefore, the power loss of the diodes can be calculated as follows

()

\subsection*{Power Loss of Capacitors}

{\label{power-loss-of-capacitors}}

Due to the power loss of capacitors is caused by the ESR of capacitor,
the power loss of the capacitors can be expressed.

()

\subsection*{Efficiency Calculation}

{\label{efficiency-calculation}}

The total losses can be calculated as

()

The efficiency calculation of the proposed converter can be calculated
as following.

()

\section*{COMPARISON}

{\label{comparison}}

Comparing performance between the proposed converter and relevant
converters is provided in Table 1. Fig. 5 (a) and (b) show the
comparison results under \emph{n} =1 with voltage gain in both step-up
and step-down modes, respectively. Fig. 6 (a) and (b) give voltage
stress on power switch, and current stress on power switch,
respectively.

In step-up mode, the proposed converter has higher voltage gain than
converter in Banaei and Ajdar Faeghi Bonab\textsuperscript{14},
Sarikhani, et al.\textsuperscript{24} , and Wang, et
al.\textsuperscript{27}. In step-down mode, converters in Banaei and
Ajdar Faeghi Bonab\textsuperscript{14}, Dhimish and
Schofield\textsuperscript{15}, Kumar and
Krishnasamy\textsuperscript{17}, and Hosseinpour, et
al.\textsuperscript{30} of voltage gain are less than proposed
converter. Although the voltage gain in Alizadeh, et
al.\textsuperscript{21} is equal to the proposed converter, the current
stress of power switch in this converter is considerably larger than the
proposed converter. In the operating condition of experiment, the
voltage stress of power switch has the superior performance comparing
with the relevant converters except for the power switch\emph{S}
\textsubscript{2} of the converter in Wang, et al.\textsuperscript{27}.
Additionally, the current stresses of power switches \emph{S}
\textsubscript{1} and \emph{S} \textsubscript{2} in Wang, et
al.\textsuperscript{27} are higher than the current stresses of\emph{S}
\textsubscript{1} and \emph{S} \textsubscript{2} in the proposed
converter, respectively. Moreover, the proposed converter has lower
number of inductors and higher frequency than relevant converters to
decrease the volume of the converter.

Table Comparing performance between the proposed converter and relevant
converters.\selectlanguage{english}
\begin{longtable}[]{@{}lllllllll@{}}
\toprule
Topology & Converter in Banaei and Ajdar Faeghi
Bonab\textsuperscript{14} & Converter in Dhimish and
Schofield\textsuperscript{15} & Converter in Kumar and
Krishnasamy\textsuperscript{17} & Converter in Alizadeh, et
al.\textsuperscript{21} & Converter in Sarikhani, et
al.\textsuperscript{24} & Converter in Wang, et al.\textsuperscript{27}
& Converter in Hosseinpour, et al.\textsuperscript{30} & Proposed
converter\tabularnewline
\midrule
\endhead
S & 1 & 1 & 1 & 1 & 2 & 2 & 2 & 2\tabularnewline
L & 3 & 2 & 3 & 3 & 3 & 2 & 3 & 2\tabularnewline
C & 4 & 6 & 5 & 3 & 3 & 2 & 4 & 3\tabularnewline
D & 2 & 4 & 3 & 5 & 2 & 2 & 3 & 3\tabularnewline
Number of Components & 10 & 13 & 12 & 12 & 10 & 8 & 12 &
10\tabularnewline
C.G & Yes & Yes & No & Yes & Yes & Yes & No & Yes\tabularnewline
Voltage Gain (\emph{M}) & & & & & & & &\tabularnewline
Voltage Stress of Switch \emph{S}\textsubscript{1}
(\emph{V\textsubscript{ds}}\textsubscript{1}/\emph{V\textsubscript{in}})
& & & & & & & &\tabularnewline
Voltage Stress of Switch \emph{S}\textsubscript{2}
(\emph{V\textsubscript{ds}}\textsubscript{2}/\emph{V\textsubscript{in}})
& - & - & - & - & & & &\tabularnewline
Current Stress of Switch \emph{S}\textsubscript{1}
(\emph{I\textsubscript{ds}}\textsubscript{1}/\emph{I\textsubscript{o}})
& & & & & & & &\tabularnewline
Current Stress of Switch \emph{S}\textsubscript{2}
(\emph{I\textsubscript{ds}}\textsubscript{2}/\emph{I\textsubscript{o}})
& - & - & - & - & & & &\tabularnewline
Frequency & 40 kHz & 75 kHz & 50 kHz & 50 kHz & 60 kHz & 100 kHz & 50
kHz & 100 kHz\tabularnewline
\bottomrule
\end{longtable}

S = Switch, D = Diode, C = Capacitor, C.G = Common ground between input
sources and load.

(a) (b)

Fig. Comparison results of voltage gain in (a) step-up and step-down
mode; (b) step-down mode.

(a) (b)

Fig. 6 Comparison of power switches on (a) voltage stress; (b) current
stress.

\section*{EXPERIMENTAL RESULTS}

{\label{experimental-results}}

To verify and test the presented converter, Table 2 demonstrates the
experimental parameters.

From (15), the curves of voltage gain versus duty cycle in various turns
ratio are plotted in Fig. 7. Considering the converter operate in the
step-up mode and step-down mode with suitable duty cycle, the turns
ratio \emph{n} is selected as 1.

The experiment waveforms under step-up mode
under\emph{V\textsubscript{in}} = 15 V, \emph{V\textsubscript{o}} = 48
V, and\emph{P\textsubscript{o}} = 100 W are illustrated in Fig. 8. Fig.
8 (a) and (b) show the voltage stresses of power switches
(\emph{V\textsubscript{ds}} \textsubscript{1}
and\emph{V\textsubscript{ds}} \textsubscript{2}) which are 31 V and 62
V, respectively, verifying (19) and (20). The voltage spike of power
switches can be inhibited by clamp circuit. The current waveform of
inductor \emph{L} \textsubscript{1} matches the expected behavior based
on theoretical analysis. Fig. 8 (c), (d) and (e) show the voltage
waveforms and current waveforms of the diodes \emph{D}
\textsubscript{1},\emph{D} \textsubscript{2}, and \emph{D}
\textsubscript{3}, respectively. The voltage stresses of
\emph{V\textsubscript{D}} \textsubscript{1},\emph{V\textsubscript{D}}
\textsubscript{2}, and\emph{V\textsubscript{D}} \textsubscript{3} are
about 30V, 30V, and 94V, respectively. Therefore, the (16), (17) and
(18) are evidenced. The current of \emph{D} \textsubscript{2} achieves
ZCS when\emph{D} \textsubscript{2} is turned off. At the same time, the
leakage inductor on the secondary side of the coupled inductor is
resonant with the parasitic capacitor of \emph{D} \textsubscript{2}.
Thus, the voltage of diode \emph{D} \textsubscript{2} have resonance
phenomena, affecting the current and voltage of
\emph{L\textsubscript{k}} .\emph{L\textsubscript{k}} resonates with the
parasitic capacitors of the diodes and power switches during diodes are
turn-on, resulting in the resonance of the voltages and currents of the
semiconductor.

The waveforms of experiment under step-down mode
under\emph{V\textsubscript{in}} = 75 V, \emph{V\textsubscript{o}} = 48
V, and\emph{P\textsubscript{o}} = 100 W are shown in Fig. 9. Fig. 9 (a)
and (b) show the voltage stresses of power switches
(\emph{V\textsubscript{ds}} \textsubscript{1}
and\emph{V\textsubscript{ds}} \textsubscript{2}) which are 103 V and 76
V, respectively, verifying (19) and (20). Similar to the step-up mode,
the voltage spike can be suppressed. Fig. 9 (c), (d) and (e) show the
voltage waveforms and current waveforms of the diodes\emph{D}
\textsubscript{1}, \emph{D} \textsubscript{2}, and\emph{D}
\textsubscript{3}, respectively. (16), (17) and (18) are evidenced with
the voltage stresses of\emph{V\textsubscript{D}}
\textsubscript{1},\emph{V\textsubscript{D}} \textsubscript{2},
and\emph{V\textsubscript{D}} \textsubscript{3} which are about 103 V,
103V, and 180 V. Considering the parasitic capacitor of diode\emph{D}
\textsubscript{2}, the voltage of \emph{D} \textsubscript{2} will
resonate with the leakage inductor on the secondary side of the coupled
inductor, also affecting the primary side current of the coupled
inductor. Therefore, the current of \emph{L\textsubscript{k}} has
resonance phenomena during the rising time. Moreover, the value
of\emph{i\textsubscript{D}} \textsubscript{2} is related to the slope of
currents of \emph{L\textsubscript{m}} and \emph{L\textsubscript{k}} . In
both step-up and step-down modes, the currents
of\emph{L\textsubscript{m}} and \emph{L\textsubscript{k}} have
significant differences so that \emph{i\textsubscript{D}}
\textsubscript{2} has different waveforms. The voltages and currents of
the semiconductor have resonance phenomena because
\emph{L\textsubscript{k}} resonates with the parasitic capacitance of
the semiconductor at the turn-off time of power switches.

The experimental results in the closed-loop for output voltage under
both step-up and step-down modes are given in Fig. 10 (a) and (b),
respectively. The load current changes from 50\% of the rated value to
100\% and then back to 50\% of the rated value. The stability
performances of the dynamic response are presented.

Fig. 11 (a) and (b) show the block diagram of power losses distributions
in step-up mode under \emph{V\textsubscript{in}} = 15
V,\emph{V\textsubscript{o}} = 48 V, and \emph{P\textsubscript{o}} = 100
W and step-down mode under \emph{V\textsubscript{in}} = 75
V,\emph{V\textsubscript{o}} = 48 V, and \emph{P\textsubscript{o}} = 100
W, respectively. Both of these modes, the diodes loss is the highest
part of the total losses. To improve the efficiency, the lower forward
voltage drops and resistance of diodes can be selected.

The curves of measured efficiency versus output power in both step-up
and step-down modes are plotted in Fig. 12. The maximum efficiency when
the converter operates in the step-up mode and step-down mode are 95.7\%
and 96.8\%, respectively. The full load efficiency of the step-up mode
and step-down mode are 86.8\% and 96.2\%, respectively.

Table the experimental parameters\selectlanguage{english}
\begin{longtable}[]{@{}ll@{}}
\toprule
Parameter & Type \& Value\tabularnewline
\midrule
\endhead
Input voltage (\emph{V\textsubscript{in}}) & 15 V\textasciitilde{}75
V\tabularnewline
Output voltage (\emph{V\textsubscript{o}}) & 48 V\tabularnewline
Rated output power (\emph{P\textsubscript{o}}) & 100 W\tabularnewline
Switching frequency (\emph{f\textsubscript{s}}) & 100 kHz\tabularnewline
Capacitors (\emph{C}\textsubscript{1}, \emph{C}\textsubscript{2},
\emph{C\textsubscript{o}}) & \emph{C}\textsubscript{1} \&
\emph{C}\textsubscript{2}: 100 \selectlanguage{greek}µ\selectlanguage{english}F \emph{C\textsubscript{o}}: 220
\selectlanguage{greek}µ\selectlanguage{english}F\tabularnewline
Power diodes (\emph{D}\textsubscript{1}, \emph{D}\textsubscript{2},
\emph{D}\textsubscript{3}) & \emph{D}\textsubscript{1} \&
\emph{D}\textsubscript{2}: MBR20300 \emph{D}\textsubscript{3}:
IDH10G65\tabularnewline
Power switches (\emph{S}\textsubscript{1}, \emph{S}\textsubscript{2}) &
IPP110N20N\tabularnewline
Inductor (\emph{L}\textsubscript{1}) & 155 \selectlanguage{greek}µ\selectlanguage{english}H Ferrite core PQ3230
Material PC95\tabularnewline
Magnetizing inductor (\emph{L\textsubscript{m}}) & 121 \selectlanguage{greek}µ\selectlanguage{english}H Ferrite core
EE4220 Material PC40\tabularnewline
Leakage inductor (\emph{L\textsubscript{k}}) & 1.69 \selectlanguage{greek}µ\selectlanguage{english}H\tabularnewline
Turns ratio (\emph{N\textsubscript{p}} : \emph{N\textsubscript{s}}) &
20:20\tabularnewline
\bottomrule
\end{longtable}

(a) (b)

Fig. The voltage gain versus the duty cycle with various turns ratio
under (a) step-up and step-down mode; (b) step-down mode.

(a) (b)

(c) (d)

(e)

Fig. Experiment waveforms under boost mode at \emph{V\textsubscript{in}}
= 15V, \emph{V\textsubscript{o}} = 48V and \emph{P\textsubscript{o}} =
100W (a) \emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{gs}}
\textsubscript{2},\emph{V\textsubscript{ds}} \textsubscript{1},
\emph{i\textsubscript{L1}} ; (b) \emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{ds}} \textsubscript{2},
\emph{i\textsubscript{Lk}} ; (c) \emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{D}} \textsubscript{1},
\emph{i\textsubscript{D1}} ; (d) \emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{D}}
\textsubscript{2},\emph{i\textsubscript{D}} \textsubscript{2};
(e)\emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{D}}
\textsubscript{3},\emph{i\textsubscript{D}} \textsubscript{3}.

(a) (b)

(c) (d)

(e)

Fig. Experiment waveforms under boost mode at \emph{V\textsubscript{in}}
= 75V,\emph{V\textsubscript{o}} = 48V and \emph{P\textsubscript{o}} =
100W (a)\emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{gs}}
\textsubscript{2},\emph{V\textsubscript{ds}} \textsubscript{1},
\emph{i\textsubscript{L1}} ; (b) \emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{ds}} \textsubscript{2},
\emph{i\textsubscript{Lk}} ; (c) \emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{D}} \textsubscript{1},
\emph{i\textsubscript{D1}} ; (d) \emph{V\textsubscript{gs}}
\textsubscript{1},\textsubscript{\emph{VD} 2}, \emph{i\textsubscript{D}}
\textsubscript{2}; (e) \emph{V\textsubscript{gs}}
\textsubscript{1},\emph{V\textsubscript{D}}
\textsubscript{3},\emph{i\textsubscript{D}} \textsubscript{3}.

(a) (b)

Fig. Closed-loop experimental results in (a) Boost mode; (b) Buck mode.

(a) (b)

Fig. 11 Block diagram of power losses distributions in (a) step-up mode
(b) step-down mode.

Fig. The curves of measured efficiency versus output power in both
step-up and step-down modes.

\section*{CONCLUISION}

{\label{concluision}}

In this article, a coupled inductor buck-boost converter based dual
switches is presented. The circuit has many advantages are as follow:
(1) combing the passive clamp circuit to suppress the generation of
power switching voltage spikes and recover the leakage energy; (2) the
high voltage gain and wide conversion ratio can be achieved by changing
the turns ratio of the coupled inductor and duty cycle; (3) common
ground between input sources and load; (4) continuous input current.
Steady-state analysis and design of components are presented in detail.
Power loss analysis and comparison with some key characteristics of
other converters are also presented and discussed. Finally, the results
of the implemented prototype verify the correctness of the theoretical
analysis, while the maximum efficiency in step-up mode and step-down
mode are 95.7\% and 96.8\%, respectively.

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